The present invention relates to power circuits. Several such circuit arrangements are described in the literature. The increase in power density and the constantly expanding hybrid structure of power circuits requires far-reaching solutions to attenuate, reduce or even eliminate parasitic effects to increase the reliability of these circuits. Great efforts are underway to find improved circuit variants to improve short-circuit resistance.
DE 4,105,155 introduces the optimization of commutation circuits of a converter power circuit. In that invention, the switchable component of the commutation circuit is positioned close to the switching semiconductor component. This at least reduces parasitic inductivities. The minimizing of parasitic inductivities reduces the overcurrents in the switching process and thus allows a higher capacitance of the circuit.
DE 4,240,501 introduces a power semiconductor circuit whose positive and negative power inputs axe divided to reduce their inductivities. The findings from a certain symmetry of the circuit in the direct current branch are being further extended. The gain in power density in the circuits when the packing density is increased is more and more influenced by the drive in all circuit elements.
In the case of densely stacked primary power and drive connections, especially circuit breakers of high switching speed and high di/dt values (fast switches), the magnetic field of the main power supplies, which is built up by every individual power bus bar, influences each drive circuit in the immediate vicinity. In that case, reactions and mutual interference can occur, which is already mentioned in EP 0 427 143 B1, without the physical processes of the transformer coupling being described. Our own earlier application (P195 38 328.1-32) had as its objective to introduce a reactionless drive of power semiconductor components without transformer effect in power circuits by means of the magnetic decoupling of the drive circuit and the main power circuit. There, the problem is solved through exact specifications for the geometric layout of the electric lines in the drive circuit.
DE 4,410,978 A1 describes a process and an associated circuit for the improvement of the short-circuit resistance of a bipolar IGBT. In that invention the circuit is provided with a MOS transistor to reduce the voltage applied to the gate electrode of the IGBT during the short-circuit state which reduces the short circuit current, which flows through the IGBT. The Zener diodes, used in this switching arrangement to realize the gate dielectric strength, limit the voltage between the MOSFET and the gate electrode of the IGBT. From a study entitled "Further development and testing of a two stage drive circuit for IGBT" by P. Nagengast, University of Erlangen, Chair for Electric Drives, dated Nov. 12, 1992, such a short-circuit protected driver circuit is known, which works with a low number of components.
From DE 4,320,021 A1 and U.S. Pat. No. 4,719,531, two switching arrangements are known which describe VCE monitors of individual switches in circuits. With such switching arrangements a limited short-circuit protection can be achieved, but such short-circuit monitors are unsuitable for parallel-connected circuit-breakers. Because V.sub.CE monitors are temperature-dependent, the breaking-current overvoltage is not increased as desired.
During the 24th Colloquium for Semiconductor Power Components on Nov. 13, 1995 in Freiburg, S. Konrad introduced a "Protection concept for voltage-controlled power-semiconductors". Here, the gate-emitter voltage of an IGBT switch is limited to a certain path through the installation-of a MOSFET.
Referring to FIG. 1, a detail of a conventional circuit arrangement with an individual switch without limiting elements for the gate drive is shown. FIG. 1 also shows a customary half-bridge circuit with IGBTs T.sub.1 and T.sub.2 and their respective driver circuits. The end phases T.sub.3 /T.sub.4 of the TOP driver and T.sub.5 /T6 of the BOTTOM driver are shown with associated output circuits. For clarification, the so-called Miller capacitances C.sub.1 and C.sub.2 are shown as external capacitances. It is well known that capacitances C.sub.1 and C.sub.2 (the Miller capacitances) are integrated in the structure of a conventional IGBT switch.
Either IGBT T.sub.1 or T.sub.2 is switched on when its respective gate is charged via top MOSFET T.sub.3 or T.sub.5 of its respective driver end phase with corresponding gate resistor R.sub.1 or R.sub.2 on +15V of the driver operating voltage source, while the serial-connected POSIT blocks T.sub.1 or T.sub.2.
Either IGBT T.sub.1 or T.sub.2 is switched off when its associated MOSFET T.sub.3 or T.sub.5 is blocked and its respective gate is discharged to -8V. Each respective gate discharges via its corresponding series-connected MOSFET (T.sub.4 or T.sub.6) and resistors (R.sub.1 and R.sub.3 or R.sub.2 and R.sub.4). As with all MOS-controlled switches, activation and deactivation is accomplished by charging and discharging the gate-emitter capacitance. Due to the voltage rise between the collector and emitter, the Miller capacitance (C.sub.1 or C.sub.2) of the IGBT (T.sub.1 or T.sub.2) must be re-charged as well.
If a short circuit occurs in the circuit of the lead IGBT, that IGBT represents the determining, current-limiting element in that circuit. The amplitude of the short-circuit current is determined by the finite amplification and the momentary gate-emitter voltage of the IGBT. The voltage rise dU.sub.CE /dt at the collector-emitter path of IGBT T.sub.1 or T.sub.2, which is very positive because of the rising short-circuit current, causes a recharging of Miller capacitance C.sub.1 or C.sub.2.
However, the recharging current pulses through the Miller capacitance (C.sub.1 or C.sub.2) increasing the gate-emitter voltage of the IGBT (T.sub.1 or T.sub.2), since its gate is decoupled by resistor R.sub.1 or R.sub.2 against driver end phase, so that the short-circuit current can rise further. This can lead to short-circuit current pulses whose amplitude may lie far above the +15V gate-emitter voltage prescribed by the driver voltage.
The less favorable the relationship between the gate-emitter capacitance and the Miller capacitance of the IGBT switch, the greater the danger of switch failure. The gate voltage of a conventional IGBT is limited by using conventional suppressor diodes. An example of limiting a gate voltage with a diode is shown in the previously discussed study entitled "Further development and testing of a two stage drive circuit for IGBT" (1993).
In this circuit, a disadvantage is the temperature dependence and the tolerance of the suppressor diodes. In the most recent generation of conventional IGBT switches, such a circuit arrangement is no longer able to provide sufficient short-circuit resistance.
FIG. 2 shows a detail of a conventional TOP driver, analogous to the TOP driver in FIG. 1, with three parallel connected power switches. As the circuit diagram indicates, the main emitters as well as the auxiliary emitters are parallel connected. If the auxiliary emitters are not parallel-connected and if only one auxiliary emitter connection is provided, oscillations may occur in the power switches. This is especially true in the case of a short-circuit. When a short circuit occurs the gate of each power switch is no longer associated with its own emitter via the drive circuit. The parallel-connection of the auxiliary emitters eliminates this association problem. However, circular (loop) currents are possible via the auxiliary emitter/main emitter circuit. Auxiliary emitter resistors (Re) are used to limit the amplitudes of these circular currents. The auxiliary emitter resistors have a preferred value of 0.5 Ohms. This means that the voltage reference for the drive circuit is the mean emitter voltage. However, the gate voltage can again be influenced through the circular currents via these resistors. This occurs in particular when there is an asymmetry in the current of the load modules.
The poor performance of the circuit shown in FIG. 2 will be investigated though a discussion of FIG. 2a and are addressed in the present invention as shown in FIG. 2b.
FIG. 2a shows the situation that occurs when two modules have already cut out the current during a commutation process and only one module still has power. The left part of FIG. 2a shows the three module currents during the turn-off process. The di/dt of current 1 becomes considerably smaller after currents 2 and 3 have already reached zero. As indicated in curves 2 and 3, the inductivities for the minus connection of the intermediate circuit are no longer effective for currents 2 and 3. This is a result of the currents 2 and 3 not being large. Since the currents are not large, they can be disregarded in the equivalent circuit diagram, but inductivity 1 is still effective.
This inductivity, which normally lies outside the gate drive circuit, now suddenly becomes effective, namely through the configuration of the auxiliary emitter resistors, in this example, with 2/3 of its value. The emitter degeneration of the slowest IGBT is once again amplified by this, and the IGBT switching transistor becomes even slower. In principle, these effects may occur with every module in a half-bridge with parallel connection.
A considerable reduction of the adverse effects is possible because the resistive voltage divider of the supplementary emitter resistors does not become effective during the cut-off phase. This can be achieved most easily through diodes parallel to the auxiliary emitter resistors. It must be taken into account that the voltage at the remaining inductivity on the side toward the interim circuit is positive during the cut-off phase. As shown in FIG. 2b, for the diode to be effective, its cathode must be connected with the emitter of the IGBT. This eliminates the strong effect of the inductivity outside the gate drive circuit. These diodes must be provided for each IGBT to ensure that all possible current asymmetries are covered by this measure. Preferably, the diodes used are Schottky diodes which have lower forward voltages.
FIG. 3 shows a comparable arrangement, according to the prior art, with a transistor insert. FIG. 3 represents a second possibility according to prior art that was already published in the above-named study in 1993. When a short circuit occurs, the gate-emitter voltage of T.sub.1 exceeds the required value of the driver operating voltage of +15V plus the value of the base-emitter path of the pnp transistor T.sub.7, the transistor T.sub.7 controls and takes over the recharge pulse of the Miller capacitance. Thus, the gate-emitter voltage of T.sub.1 is limited.
The solution in FIG. 3 can not be used in the case of parallel-connected IGBTs because the operating voltage, as a voltage reference, can be clearly assigned to the emitters of each individual IGBT but refers to the mean value of the emitter potentials. Another solution must be found for parallel connected IGBTs.
FIG. 4 shows a gate voltage limiting device according to the prior art, where a Schottky diode (D.sub.3) is used with a half bridge circuit without parallel-connection. A limiting circuit is shown with only one additional component per IGBT switch. A fast diode D.sub.3 bridges resistor R.sub.1 in case of a short-circuit in the on-phase and leaks the recharge current pulse of the corresponding Miller capacitance to the driver supply voltage source via the antiparallel diode integrated in the T.sub.3 MOSFET.
The effectiveness of this circuit is detrimentally affected by the parasitic inductiveness of the connection lines between the driver end phase and the gate and emitter ports of the IGBT. This is a disadvantage that does not occur in all design variants of circuit arrangements. The D.sub.3 diodes and the diodes integrated in the T.sub.3 transistor act as limiting diodes.
FIG. 5 shows a conventional gate-emitter voltage limiter by means of an additional Schottky diode D.sub.3 and a capacitor C.sub.3 in the immediate vicinity of the gate-emitter ports of IGBT T.sub.1. Once again, the figure sketches a detail from a circuit of the driver end phase of the TOP driver.
The capacitor C.sub.3 is charged via the resistor R.sub.5 (with a typical value of about 10 Ohm) to the original +15 V of the driver. An adequate gate-emitter voltage limitation is achieved when capacitance C.sub.3 is clearly greater than the gate-emitter capacitance of the IGBT in question. In the case of a short circuit, capacitor C.sub.3 can absorb the recharge current pulse through the Miller capacitance, without much rise in the voltage.
FIG. 6 shows a conventional gate drive circuit for parallel-connected half-bridges analogous to FIG. 5. This circuit is less suitable for a parallel circuit because the emitter inductivities can cause interaction with the capacitors in the form of oscillations, especially true in the case of asymmetric current patterns.
FIG. 7 shows a variant of a conventional circuit arrangement, again showing the detail for a TOP driver, with a reference diode. By using a temperature-stabilized voltage reference diode D.sub.7 in combination with a pnp transistor T.sub.7, a solution can be achieved that is analogous to the prior art shown in FIG. 3. This circuit variant has a disadvantage in that the base-emitter path of the pnp transistor T.sub.7,with the IGBT turned off, must block T.sub.1 at about 8V. This problem can be minimized by reducing the negative supply voltage of the driver from -8V to -5V. This is not a practical limitation in most cases.